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Aug 16 2019

Design of Numerical Control DC Adjustable Switching Power Supply

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Switching Power Supply

 

 

 

 

 

 

 

 

 

 

 

 

 

 

1. Design of Hardware Circuit

1.1 Main Power Conversion Circuit

1.1.1 Design of 220V Input Anti-interference and Rectification Filter Circuit


1.1.2 Design of Main Power Switching Circuit


1.1.3 Design of PWM Control and Voltage-regulated Circuit


 

 

1.1.4 Design of Transformer

1.1.4.1 Design of Main Power Transformer

1.1.4.2 Design of Excitation Transformer



1.2 Design of Output Circuit

1.2.1 Design of BUCK Synchronous Step-down Rectifier Circuit


1.2.2 Drive Circuit


1.2.3 Feedback Sampling Circuit


 

1.3 Design of Main Control and Display Module

1.3.1 Minimal System and Key Circuit Design of Single Chip Microcomputer


1.3.2 Design of Liquid Crystal Display Circuit


1.4 Design of Auxiliary Power Supply

1.4.1 Design of Auxiliary Power Supply Circuit


1.4.2 Design of Auxiliary Power Transformer







2. System Debugging

 

 

 2.1 Hardware Debugging

2.1.1 Main Power Conversion Circuit Debugging


2.1.2 Output Circuit Debugging


2.1.3 Auxiliary Circuit Debugging


2.2 Machine Test



2.3 Physical Map of Various Modules of the System




1. Design of Hardware Circuit

The hardware circuit is composed of a main power conversion circuit, an output circuit, a main control and display circuit, and an auxiliary power supply circuit. The main power conversion circuit includes a 220V input rectification circuit, a switching circuit, an output rectification and filtering circuit, and a PWM circuit. The output circuit includes a BUCK synchronous buck circuit, a driving circuit, a filtering circuit, and a feedback circuit. The main control circuit includes the STM32 minimum system circuit, and the LCD12864 display circuit. The auxiliary power circuit includes a self-excited switching power supply circuit and a circuit which turns from 5V to 3V.

1.1 Main Power Conversion Circuit

The main power conversion circuit converts the industrial frequency AC power of 220V and 50HZ into 50V DC power. This part adopts the isolated half-bridge switching power supply topology, and uses TL494 to generate PWM wave to drive the excitation transformer to drive the switch to turn on and off. The error amplifier inside the TL494 makes the voltage regulated and the current constant to realize voltage conversion.

1.1.1 Design of 220V Input Anti-interference and Rectification Filter Circuit

The schematic diagram of 220V input rectification and filtering is shown in figure 1. The 220V AC input to the system first passes through a two-stage EMI filter circuit. The two stages of EMI filter circuit are composed of C32, L4, C33, C28, L5, C27, C34, which is essentially a low-pass filter circuit, allowing 50Hz industrial frequency AC power to pass through, thereby suppressing the influence of high-frequency harmonics in the power grid on the entire system, and also preventing high-frequency harmonics generated by the system itself from entering the power grid and interfering with other power-consuming equipment.

After the 220V AC of the EMI filter circuit passes through the passive PFC circuit for power factor correction, the passive PFC circuit is a low-pass filter circuit composed of C30, R33, L3. The inductor L3 acts as a buffer compensation here, reducing the phase difference between the voltage and current of the input alternating current, thereby increasing the power factor of the system.

After the calibration, the 220V AC power is full-bridge rectified by the full bridge ZL1, and then filtered by C29 and C35 to obtain a stable DC voltage, and C29 and C35 are charged to about 160V, and are supplied to the half bridge switching circuit.

 Figure 1. 220V Input Rectification and Filtering Circuit Diagram

Figure 1. 220V Input Rectification and Filtering Circuit Diagram

1.1.2 Design of Main Power Switching Circuit

The main power switch circuit adopts a half bridge switching topology, and the circuit schematic is shown in figure 2. The two switching tubes Q1 and Q2 are alternately turned on in one switching cycle, and the PWM signals from the pin 9 and 11 of the TL494 are given to the base electrode of the push-pull driving tubes Q3 and Q4 to make them alternately turned on. The 12V DC power supplied by the auxiliary power supply passes through the primary windings N1 and N2 of T2 to the collectors of Q3 and Q4, and the primary of T2 generates an alternating voltage. The secondary windings N3 and N4 of T2 respectively induce opposite voltages, which are respectively applied to the base electrode of Q1 (MJE13009) via D2, R15, and R16, and are added to the base electrode of Q2 (MJE13009) via D5, R20, and R21 to make it alternately turn on. When Q1 is turned on, the voltage on C29 passes through Q1, secondary winding N5 of T2, primary winding N6 of T1 and C10 to form a discharge loop. At this time, the voltage on the primary winding N6 of T1 is from bottom to top. When Q2 is turned on, the voltage on C35 passes through C10, the primary winding N6 of T1, the primary windings N5 of T2 and Q2 to form a discharge loop. At this time, the voltage on the primary winding N6 of T1 is from top to bottom. N7 and N8 induce voltage by the alternating current on N6 and the voltage is fully rectified by D3 and D4 to obtain 50V voltage. R14, C11 and R23 and C16 form an absorption circuit to reduce the peak voltage generated when the diode is turned on and off. The rectified voltage passes through L1, C13, L2, and C12 to form a wave filtering of a second-order LC filter circuit. To prevent the output voltage from being false high, the R19 pseudo load is applied to stabilize the voltage.

In order to reduce the switching losses of Q1 and Q2, that is, to reduce the time for the transistors Q1 and Q2 to enter the saturation region from the amplification region, the design incorporates an accelerating circuit composed of C9, D2, R15 and C14, D5, R20 and N5. When Q1 is turned on, N5 flows through the current from left to right, and N3 has the same phase, which will induce a right-to-left current. After D2 and R15, the forward bias current increases, and the saturation depth is increased. The acceleration of C9 (the voltage across the capacitor is instantaneously constant) causes Q1 to quickly enter the saturation region from the amplification region. Q2 is the same.

 Figure 2. Main Power Switching Circuit Diagram

Figure 2. Main Power Switching Circuit Diagram

1.1.3 Design of PWM Control and Voltage-regulated Circuit

The PWM control circuit uses TL494 as the core control period. The TL494 is a well-functioning voltage-driven pulse width debugging device produced by Texas Instruments. It can be used as a control device for single-ended, push-pull and bridge switching power supplies. It can be designed as single-ended output and double-ended push-pull output. This design uses double-ended push-pull output and its internal block diagram is shown in figure 3.

 Figure 3. Internal Block Diagram of TL494

Figure 3. Internal Block Diagram of TL494

The TL494 integrates two error amplifiers, which are used as voltage feedback and current feedback in this design. The values of voltage and current are set to the inverting input of the error amplifier, and the feedback value is applied to the non-inverting input of the error amplifier. Here, the error amplifier is regarded as an ideal op amp. It is known from the characteristics of the op amp that if the feedback voltage or current exceeds the set value, a high level is output to make the PWM comparator work and reduce the duty ratio of PWM output, thereby reducing the output voltage and current. The oscillation frequency is set by the pin 5 and 6, and the oscillation frequency is realized by an external resistor and capacitor. Pin 4 is the dead time control, which can limit the minimum output dead time to 4% of the PWM cycle. When it is connected to the low level, the output duty ratio is 96%. When it is connected to the high level, the output duty ratio is 48%.

Figure 4 is the PWM control circuit of the main power switching circuit. R13 is 18K, C8 is 1000PF, and the calculated switching frequency is about 61KHz. The set voltage and current are supplied by an internal 5V reference voltage, and the appropriate voltage value is adjusted by two potentiometers. The voltage feedback is composed of R1 and R2 and the sampling ratio is 1/10. The current feedback resistor R3 is taken with constantan wire of 0.01 ohm, and R1 is the set current limiting potentiometer. The divided voltage is input to the current error amplifier after 1/48 voltage division by R8 and R9, which is equivalent to amplifying the sampling current by 48 times, so that the feedback control is stable. Since the voltage feedback sampling ratio is 1/10 and the sampling voltage is 5V at the maximum, the voltage error amplifier input can be directly input by voltage R2. C6 is added in order to prevent the power tube with load from being burned because the instantaneous current is too large. When the power is turned on, the capacitor is charged. The instantaneous voltage of pin 4 is 5V. At this time, the duty ratio is 48%, and then the capacitor is discharged through R12 to lower the voltage to 0V, which realizes boot buffer. In order to prevent the switch from self-exciting, R4, R7, C3, and C4 are added to the PWM compensation terminal. Pin 9 and 11 are PWM output terminals that control the turn-on and turn-off of the switching tube.

 Figure 4. PWM Control Circuit

Figure 4. PWM Control Circuit

1.1.4 Design of Transformer

1.1.4.1 Design of Main Power Transformer

The transformer design includes a main power transformer and a push transformer.

The output voltage of the main power loop design is 0-40V. Considering the voltage drop and margin problem, the maximum output of the transformer design is about 50V. First, we need to select the transformer core. The following empirical formula exists between the maximum power PM (unit: W) of the high-frequency transformer and the effective cross-sectional area Sa of the core (unit: cm2).

 formula 1

The maximum output power of this design is 400W. Considering that the efficiency is about 85%, the transformer design power should be 470W or more, which is about 500W. Calculated by the formula, we can know that Sa=2.46cm2. We can select the core EE42A and its effective cross-sectional area is 1.24*1.97=2.44cm2, which basically meets the requirements.

Since the half-bridge switching power supply is also a double-excited switching power supply in principle, the magnetic flux density B of the transformer core for the half-bridge switching power supply can be from -Bm to +Bm, and the difference is that  the voltage applied to the primary winding of the transformer by the half-bridge switching power supply is one-half of the supply voltage. Therefore, the calculation formula of the half-bridge switching power supply can be derived from the calculation formula of the double-excited switching power supply. The formula for calculating the primary winding of a push-pull switching power supply transformer is:

formula 2—— Calculation of primary winding of double-excited transformer

The voltage applied to the primary winding of the transformer in the half-bridge switching power supply is 1/2 of the power supply voltage. It is set to Uab here, so the above formula used as the half-bridge switching power supply should be changed as:

formula 3—— Calculation of primary winding of half bridge transformer 

Among the formula, Ui is the input voltage (unit: V), S is the effective cross-sectional area of the core (unit: cm2), Bm is the maximum allowable flux density (unit: G), and τ is the time constant (unit: second) ), that is, the turn-on time of the switching tube in one switching cycle. Then τ=Ton=D*T=D*, where D is the duty ratio and F is the switching frequency. Considering that the grid voltage will fluctuate around 220V±10%, the corresponding input voltage is 280V-342V after full-wave rectification, and is calculated by the maximum duty ratio and the output voltage at the lowest input output voltage when calculating the maximum output voltage. Bm is generally limited to 2000G and here Bm takes 1200G. Since the maximum duty ratio of the half-bridge switching power supply is D=0.5, the calculation formula is finally derived as follows:

 formula 4

Substituting into the formula and calculating, we can get that N1 = 38 circles. Since the output voltage Uo of the half-bridge switching power supply is determined by the forward voltage of the secondary winding of the transformer, the maximum output voltage is calculated by taking the duty ratio 0.5. In a cycle, it is a square wave that is complementary to each other, and because the square wave's waveform coefficient (the ratio of the effective value to the half-wave average) is 1, its half-wave rms value is equal to the half-wave average value, so here the ratio of transformer primary to the secondary is:

 formula 5

We can get N2=14 circles.

Due to the high power of the transformer, the N1 is wound with two enamelled wires with a diameter of 1.0 mm. The N2 uses four 0.51 mm enameled wires and is wound around two groups, that is, eight enameled wires. The winding method uses a sandwich winding method, that is, the primary winding and secondary winding are stacked and wound, which helps to reduce the leakage inductance of the transformer and improve the efficiency of the transformer.

1.1.4.2 Design of Excitation Transformer

The function of the excitation transformer is to provide excitation voltage to the two power tubes of the half bridge, and to isolate the primary high voltage, so the power is small. Here, the magnetic core EE19 is used, and the effective sectional area of the magnetic core is 0.22 cm2. Since the power tube uses a triode, the Vbe only needs 0.7V or more to turn it on, so its secondary output voltage is low and about 2V. Since it is still a double-excited push-pull structure, the calculation formula of the double-excited transformer can still be used. The formula is as follows:

 formula 6

At this time, the corresponding input voltage is 12V, the frequency is 61KHz, the duty ratio is 0.5, Bm is still 1200G, and after calculation we can get N3=30circles, then the secondary N4 is 5 circles. The winding method sill uses a sandwich winding method.

1.2 Design of Output Circuit

In order to improve the efficiency of the whole machine, the output circuit adopts BUCK synchronous step-down rectifier circuit, using STM32F103C8T6 single-chip microcomputer as the controller, setting the switching frequency to 50KHz, and controlling the output voltage level by controlling the turn-on time of the upper and lower bridge arms in one cycle.

1.2.1 Design of BUCK Synchronous Step-down Rectifier Circuit

The difference between the BUCK synchronous step-down circuit and the traditional BUCK is only to replace the diode with the MOS tube, so the basic topology is the same. Figure 5 shows the BUCK synchronous step-down circuit. The input part adopts a π-type filter circuit. In order to reduce the ripple, the output circuit uses multiple capacitors for parallel filtering. In order to reduce the transient spike voltage generated when the MOS transistor is turned on and off, a 10K resistor is added between G and S of the MOS transistor to absorb the transient voltage when the MOS transistor is turned on so as to increase the resistance value of the drive resistor of the MOS tube and reduce the slope of the drive voltage and the switching noise. In order to ensure that the output voltage does not drop when carrying a large current, the energy storage inductor L7 uses a 100H/10A inductor.

 Figure 5.BUCK Synchronous Step-down Rectifier Circuit

Figure 5. BUCK Synchronous Step-down Rectifier Circuit

1.2.2 Drive Circuit

Since the switching tube of the output circuit is a MOS tube, the driving voltage is generally 10V or more and the driving force is insufficient by using the single-chip microcomputer to directly drive, the driving circuit adopts the IR2110 as the core. IR2110, developed and put on the market by American International Rectifier Company in 1990, is a high-power dedicated gate drive integrated circuit for MOS and IGBT. It has been widely used in power drive fields such as power conversion and motor speed regulation. The chip is small in size, high in integration, and the propagation time of the rising and falling edges is less than 10 ns. Its driving ability is strong, and the undervoltage lockout is built-in, which is easy to debug. In particular, the upper tube driver uses an external bootstrap capacitor to power up, so that the number of driving power channels is greatly reduced compared to other IC drivers. It only need a 10-20V power supply. Therefore, it greatly reduces the size and the number of power supply of the control transformer. It also reduces the product cost and improves system reliability. The drive circuit is shown in figure 6

 Figure 6. MOS Drive Circuit

Figure 6. MOS Drive Circuit

HIN and LIN are the high-end and low-end PWM signal inputs respectively. The PWMA and PWMB signals are output from IO of the single-chip microcomputer and the driving ability is enhanced through IR2110 so as to drive the upper and lower arms of the MOS tubes respectively. Vs is the high-end floating power supply offset voltage, Vb is the high-end floating power supply voltage, and its floating terminal is powered by the bootstrap capacitor, so that the high-end working voltage can reach 500V. In order to suppress the peak voltage of the MOS at turn-on and turn-off moment, the value of the gate drive resistors R40 and R44 should be slightly larger such as 20Ω, and D11, D13 are connected in parallel to suppress the influence of the surge current on the IR2110.

1.2.3 Feedback Sampling Circuit

Figure 7 shows the feedback sampling circuit. The feedback circuit is divided into voltage feedback and current feedback. The voltage feedback uses voltage division sampling of 10K and 150K resistors , so that the sampling voltage is 1/16, and the current sampling uses constantan wire of 0.01Ω as the sampling resistor. Because the temperature drift of the constantan wire is very small, the accuracy of the sampling is guaranteed. Since the output voltage is 0-40V, and the internal ADC range of the microcontroller is 0-3.3V, 1/16 voltage division sampling is adopted. In this way, the sampling voltage does not overflow when the output voltage is the highest. With a 150K resistor in combination with a 20K potentiometer, the potentiometer is adjusted to make the sampling more accurate. Since the output current is 10A at the maximum, the current loss will be very large if the sampling resistance is too large. To ensure the sampling accuracy, we choose 0.01 Ω constantan wire which is with very small temperature drift so that the sampling voltage is 0.1V at a current of up to 10A. Direct sampling will definitely cause a large error when the amount of change is small, so the operating voltage of the constantan wire is 20 times amplified by an operational amplifier. LM358 is a non-inverting amplifier circuit, ref=1+(R39+R38)/R43. Adjust the value of the sliding rheostat R39 during debugging so that the magnification is exactly 20 times.

 Figure 7. Feedback Sampling Circuit

Figure 7. Feedback Sampling Circuit

1.3 Design of Main Control and Display Module

1.3.1 Minimal System and Key Circuit Design of Single Chip Microcomputer

The main control circuit is shown in figure 8. The core of the main control circuit is STM32F103C8T6 single-chip microcomputer. The periphery consists of crystal oscillator circuit, reset circuit, mode selection circuit and key circuit. The crystal oscillator circuit provides a stable working pulse for the single-chip microcomputer. P1 selects the normal working mode and the download debugging mode. The four buttons respectively implement addition and subtraction of current and voltage.

The main control circuit is shown in Figure 8. The core of the main control circuit is STM32F103C8T6 single-chip microcomputer. The periphery consists of crystal oscillator circuit, reset circuit, mode selection circuit and button circuit. The crystal oscillator circuit provides a stable working pulse for the single-chip microcomputer. P1 selects the normal working mode and the download debugging mode. The four buttons respectively implement current addition and subtraction, voltage addition and subtraction.

 Figure 8. Main Control Circuit

Figure 8. Main Control Circuit

1.3.2 Design of Liquid Crystal Display Circuit

The liquid crystal display module uses LC12864 liquid crystal. The liquid crystal with Chinese character library is 5V drive and the backlight is adjustable. It is mainly composed of row driver, column driver and 128*64 full dot matrix liquid crystal display. The external interface is serial and parallel. In order to simplify wiring, this design uses serial wiring. LCD12864 and microcontroller interface circuit are shown in figure 9.

 Figure 9. Liquid Crystal Display Circuit

Figure 9. Liquid Crystal Display Circuit

1.4 Design of Auxiliary Power Supply

1.4.1 Design of Auxiliary Power Supply Circuit

The auxiliary power supply is designed to provide a stable DC voltage of 12V, 5V, and 3.3V to the control circuit. The auxiliary power supply adopts a single-tube self-excited switching power supply. Since the single-tube self-excited switching power supply component is small in amount and in size, it is preferable as an auxiliary power supply with low power consumption.

As shown in figure 10, the auxiliary power supply circuit diagram shows that the 310V DC current obtained through the input rectification and filtering circuit passes through the N1 winding of the transformer T3 to the collector of the switching transistor Q7, and simultaneously passes through the starting resistor R45 to the base of the driving transistor Q7 to provide initial voltage for the conduction of Q7. Q7 changes from the off state to the micro-conduction state, so that the primary winding N1 of the transformer generates a positive and negative voltage. Since N2 and N1 are wound in the same direction, the induced voltage of N2 is the same as that of N1. It is applied to the base of Q7 via R48, D18, C59, so that Q7 is fully turned on. At this time, the current in N1 and N2 reaches the maximum value. It is known from the law of electromagnetic induction that the voltage induced by N2 is generated by the increased current in N1. The voltage value of N2 depends on the rate of change of the current in N1. When the growth rate of N1 is slowed down, the voltage of N2 drops, so that the base voltage of Q7 decreases, the conduction level decreases, and the current in N1 begins to decrease. At this time, the upper voltage on N1 changes negative and the lower one becomes negative, and the voltage on N2 also has the same change. This voltage is connected in series with C59 and then applied to the base of Q7, so that Q7 changes from saturated state to off state. When it is turned off, C59 begins to discharge. After the discharge is completed, a new round of cycle is started under the action of the starting resistor R45. Therefore, the capacity of the C59 determines the speed of charge and discharge, and further determines the switching frequency.

The flyback power supply generates a high anti-peak voltage at the moment when the switch is turned off. To prevent it from breaking through the switch tube, D15, C56, and R46 are used here to form an absorption circuit. R51 is here to prevent the base potential of Q7 from being too high. R55 is used for current protection. When the current on Q7 is too large, the voltage on R55 rises and acts on the base of Q8 after passing through D19. After Q8 is turned on, the base voltage of Q7 decreases, which reduces the conduction level to achieve the effect of current limiting.

Figure 10. Auxiliary Power Supply Circuit 

Figure 10. Auxiliary Power Supply Circuit

The voltage closed-loop voltage-regulated circuit is composed of TL431, optocoupler, Q8 and other components. The voltage induced on the winding N4 is rectified by D16 and then filtered by a π-type filter circuit composed of C60, L9 and C61 to obtain DC voltage. The output voltage is divided by R54 and R57 and then applied to the reference terminal of TL431 to control the conduction degree of TL431. The higher the voltage is, the higher the conduction degree is. When the output voltage is too high, the equivalent resistance of TL431 resistor becomes smaller. The degree of illuminance of the coupling is enhanced, so that the conduction degree of the other side of the optocoupler is enhanced. Then the voltage on N2 is rectified and filtered by D17 and C62, and passes through the optocoupler and R52 to the base of Q8, controls the conduction degree of Q8 and further controls the conduction degree of Q7 to achieve closed-loop regulation thus stabilize the output voltage at the design value. The voltage output from the N3 and N4 windings is filtered by a half-wave rectification and a π-type filter circuit to obtain a voltage of 12V and 5V. The voltage of 3.3V is obtained by stepping down the voltage of 5V by using the LDO linear regulator LM1117-3.3.

The voltage feedback uses the TL431 produced by Texas Instruments as a feedback control component. The TL431 is a controllable precision regulator source, and its internal structure is shown in figure 11. The application circuit is shown in figure 12. The output voltage can be set to any value from Vref (2.5V) to 36V with only two resistors. There are a wide range of applications in switching power supplies.

Figure 11. Internal Structure of TL431 

Figure 11. Internal Structure of TL431

Among them, Vref is the reference of 2.5V, Vo=(1+R1/R2)*Vref. This design is to ensure that the output voltage is slightly larger than 5V, the feedback resistor R1 is 1.15K, and R2 is 1K.

1.4.2 Design of Auxiliary Power Transformer

The auxiliary power supply uses a single-tube self-excited flyback switching power supply whose switching frequency is determined by LN, the inductance of the N2 winding and C59. That is, the junction capacitance of the switching tube and the LC resonance frequency of the feedback coil N2. The switching frequency of this design is designed to be about 50KHz, and the calculation formula is as follows:

formula 7 

The magnetic core takes the core of EE19, the switching frequency is designed to be 50KHz, and the LN is 4.05uH, and take LN as 4.1uH. Next, determine the number of circles of the feedback winding. The feedback winding takes an enameled wire with a diameter of 0.31 mm. After winding 20 turns, the measured inductance is about 2.7 uH, and the measured inductance is about 4.1 after 25 turns, so take the feedback winding N2 as 25 turns.

The effective value in the closed magnetic circuit can be calculated from the inductance LN, the number of the circles of feedback coil N2, the effective magnetic path length Le of the magnetic core, the effective cross-sectional area Ae of the magnetic core, and the vacuum permeability μ0 (4π*10-7H/m). The magnetic permeability μe is calculated as follows:

 formula 8

The effective magnetic path length of the EE19 core Le=0.39 cm, and the effective sectional area Ae=0.23 cm2. So μe = 886.2.

Output voltage of designed self-excited switching power supply is 100-242V, the output voltage is 5V and the output current is 1A.

Calculate the primary current if you want to calculate the inductance of the primary winding of the self-excited switching power supply. The primary peak current is calculated by using the following formula:

 formula 9

V0 is the output voltage of 5V, I0(max) is the maximum output current of 1A, η is the efficiency of the switching power supply of 70%, Dmax is the maximum duty cycle of 0.5, and DC(min) is the minimum input voltage of 100V. The calculated Ippk is 0.286A.

The formula for the primary primary inductance calculated from the primary peak current is as follows:

 formula 10

In the formula, fs is the switching frequency, where 50KHz is taken, and Lp=3.5mH is calculated.

It is known that the effective permeability μe=886.2. The number of circles of the transformer primary coil N1 calculated from the formula is 197.8 and we take it as 200.

The formula for secondary winding calculation is:

formula 11 

Then the 5V secondary winding N3=20 circles, and the number of circles of 12V N4 is 45 circles.

The transformer winding method still uses the sandwich winding method. The primary coil is divided into two groups of 100 turns, and the innermost and outermost 100 turns of the feedback winding and the output winding are wrapped inside.

 

2 System Debugging

2.1 Hardware Debugging

2.1.1 Main Power Conversion Circuit Debugging

Since the main power circuit is in a separate working state, the debugging process can be separated from the subsequent output circuit, and the installation can be performed after the debugging is normal. Since the main power circuit directly takes power from AC, in order to ensure safety, the string bulb is used for debugging. The advantage of this debugging method is that if the switching power supply works abnormally, the current will surge, the bulb will emit light, and the luminous bulb resistance will increase greatly, and the divided voltage will increase, thereby greatly reducing the input voltage of the switching power supply, and reliably protecting the switch tube from burning and causing an explosion. After the measured output voltage is normal and there is no abnormality in operation, it is directly powered on.

We need to test whether the TL494 driver is normal before starting. Use external 12V DC voltage to the pin of power supply TL494. The measured output waveforms of the pin 9 and pin 11 is shown in figure 12. Since the feedback voltage and current are both 0, the duty ratio is maximum at this time, which is about 96%, indicating that the TL494 is working normally.

 Figure 12. Full Duty Cycle Waveform of TL494

Figure 12. Full Duty Cycle Waveform of TL494

When the string bulb starts, the bulb filament is slightly yellow, indicating that the main power switch circuit is not abnormal. Use a multimeter to measure the output voltage and get the result as 28V, adjust the feedback resistor knob, and the output voltage rises up to 54V, indicating that the feedback regulator loop works normally. Adjust the voltage to 20V with a 5 ohm power load. At this time, the voltage drops to 12.5V, indicating that the current feedback loop starts to work. The current is limited to the set value, and the current-limited potentiometer knob is adjusted, and the voltage can drop with the rise. From then on, it is determined that the voltage regulator loop and the current limiting loop are working properly.

When it works with or without load, move your ear to be close to the switching power supply, and no self-excited sound is heard, indicating that the design and the control loop of the switching power supply transformer are better handled.

After the main power loop is working normally, it is tested with load and output ripple. The load is 4 5 ohm 100W power resistors. After two parallel connections in series, the actual resistance is 4.9 ohms. The test form is shown in table 5.1.

                              Table 1. Test Table of Main Power Switch Circuit     

Number of measurements

1

2

3

4

5

6

Voltage

(V)

5.03

10.23

15.04

20.00

30.05

40.07

Current

(A)

1.027

2.088

3.070

4.082

6.125

8.172

Ripple

(mV)

31

56

74

98

124

153

As shown by the test, the main power switch circuit design meets the design requirements.

2.1.2 Output Circuit Debugging

For the output circuit debugging, firstly, the PWM output debugging of the single chip microcomputer should be carried out, then the driving circuit debugging should be carried out, and finally the output circuit debugging can be put into normal operation. The MCU first performs simple program debugging, so that it normally outputs two square complementary waves with a duty cycle of 50%. Connect it to the driver circuit IR2110, use the oscilloscope to view the output waveform of the driver circuit IR2110, and find that the high-end and low-end outputs of the driver circuit are complementary square waves with a duty cycle of 50%, and the amplitude is 12V. The measured waveform is shown in figures 13 and 14.

 Figure 13. 50% PWM Output Waveform of the Microcontroller

Figure 13. 50% PWM Output Waveform of the Microcontroller

 Figure 14. 50% PWM Drive Waveform of IR2110

Figure 14. 50% PWM Drive Waveform of IR2110

It can be seen that the MCU output and drive circuit of IR2110 are normal, and the program is programmed. Turn on the MCU and make the input voltage be 0V, and the oscilloscope is used to measure the PWM output of the MCU and the IR2110 output waveform. The waveform shown in figures 15 and 16 is obtained.

 Figure 15. PWM Output Waveform of Microcontroller at 0V Input

Figure 15. PWM Output Waveform of Microcontroller at 0V Input

Figure 16. Driving Waveform Diagram of IR2110 at 0V input 

Figure 16. Driving Waveform Diagram of IR2110 at 0V input

The program sets the PWM wave to output at the maximum duty cycle when the feedback sampling voltage and current is 0, and the limiting waveform is processed to prevent the duty cycle from being too high, so the measured waveform is basically normal, indicating that the feedback closed loop has started normally.

Connected to the input voltage of 24V, the button adjusts the voltage, and it is found that the duty cycle changes and the output voltage changes accordingly. It can be seen that the output circuit is debugged normally.

Load and ripple tests were performed with a 4.9 ohm load. The test results are shown in table 5.2.

                         Table 2. Table of Output Circuit Test with Load

Number of Measurements

1

2

3

4

5

6

Voltage (V)

5.12

10.34

15.50

20.00

30.10

40.01

Current (A)

1.10

2.10

3.24

4.12

6.15

8.17

Ripple (mV)

78

85

89

98

114

150

It can be seen that the voltage stability value and the load carrying capacity are very good, but the output peak is not well suppressed due to the turn-on and turn-off of the switch tube, so that the output ripple is greater than 100 mv in the case of high current with load. Overall, it basically meets the design requirements.

2.1.3 Auxiliary Circuit Debugging

Since the auxiliary circuit is also a switching power supply, it also needs to obtain energy from AC, so its startup is also started by a series of bulbs. When it is powered on, if you find that the bulb does not emit light, then it indicates that the auxiliary power supply is not short-circuited or self-excited. Using a multimeter to measure the output voltage 5V port and know that the measured voltage is 5.12V, and 12V port voltage is 12.5V, which is in line with design requirements. Remove the bulb, start it directly, and measure the load capacity with a 30 ohm power resistor. The test results are shown in table 5.3.

                                 Table 5.3 Auxiliary Circuit Debugging

Measuring port

5V

12V

Voltage (V)

5.12

12.32

Current (A)

0.175

0.413

It has been found that the voltage and current are in line with the design requirements.

2.2 Machine Test

The whole machine test includes output power, measurement display accuracy, output ripple, and overall machine efficiency test. The load test still uses 4.9Ω power resistor, and the data of each test are recorded separately. The test results are shown in table 5.4.

                                                  Table 5.4 Table of System Overall Test


1

2

3

4

5

6

Actual Voltage - U(V)

5.12

10.54

20.55

25.42

30.10

40.03

Display voltage - U(V)

5.10

10.60

20.60

25.40

30.10

40.00

Actual current - I(A)

1.05

2.15

4.20

5.19

6.14

8.16

Display current - I(A)

1.06

2.15

4.23

5.21

6.17

8.17

Ripple (mv)

78

85

89

98

114

150

Output Power - Pout(W)

5.43

22.66

86.31

131.93

184.82

326.65

Input Power - Pin(W)

10.63

27.82

101.66

153.86

219.56

398.75

Effectiveness

52%

81%

82%

85%

84%

82%

It can be seen from the measurement results that the voltage and current display accuracy meets the design requirements, the ripple is within a reasonable range, and the current limit and the voltage regulation are normally available during the adjustment. Since the transformers are all wound by themselves, the core gap control may be poor, the leakage inductance may be relatively large, and the component selection and the inherent loss are caused, resulting in a no-load power consumption of 5.2W, so the efficiency is not ideal when it is with a small power load. According to other data tested, the load effect is better in the case of high power load, and the efficiency meets the design requirements.

2.3 Physical Map of Various Modules of the System

The physical map of the system is shown in figure 17. It consists of auxiliary power module, main power switch circuit module, LCD12864 display module, main control module, BUCK synchronous buck module and power input.

Figure 17. Physical Map of the System 

Figure 17. Physical Map of the System

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